Test of automatic calibration systems for room characteristics (Room EQ). Principles of frequency response correction. Purpose of correction circuits Adjustment of frequency response

Broadcast channel links introduce amplitude-frequency distortions. This means that their gain or attenuation is a function of frequency and the frequency response of the gain is not horizontal.

In many broadcast devices, the magnitude of amplitude-frequency distortions, manifested as a decrease in the transmission coefficient at extreme frequencies, is reduced to a normalized value by rational construction electrical diagram, the choice of the values ​​of its elements and operating mode, the use of negative feedback. But the amplitude-frequency characteristics of some parts of the broadcast channel, connecting lines, sound recording and sound reproduction devices, long-distance lines, and wire broadcasting lines do not have a horizontal section. In these cases, amplitude-frequency distortions are reduced by including a special circuit in the broadcast channel - correction contour CC.

Adjustment principles

Amplitude-frequency response QC should be such that the overall amplitude-frequency characteristic of the distorting link and. QC in a given frequency band from fmax to fmin was a horizontal line. So, the condition for frequency correction of the distorting link:

where and - respectively, the attenuation (transmission) coefficient of the distorting link and the correction circuit.

Methods for correcting amplitude-frequency distortions in terms of technical methods and calculation methods are close to methods of frequency predistortion. Frequency pre-emphasis is the artificial distortion of the spectrum of a broadcast signal in order to improve the SNR. Frequency pre-emphasis is widely used in broadcast channels, such as trunk lines, sound recording devices, and frequency modulation radio broadcasting.

Since trunk lines are included in the broadcast channel in various arbitrary combinations, they are considered as independent links of the channel. It is undesirable to compensate for amplitude-frequency distortions introduced by the trunk line in other links of the channel - LU or PU, since in that case it is impossible to maneuver the amplifiers and trunk line and connect any trunk line to any amplifier. Each SL must be adjusted, regardless of other links in the channel. The identity of the frequency response of the corrected lines facilitates their operation and mutual redundancy. The frequency response of the corrected SL must fit within the template:

SL uses fundamentally different methods for adjusting the frequency response than in wired broadcasting lines. Due to the large number of trunk lines sequentially included in the broadcast channel, high accuracy of correction is required (see Table 1).

The connecting lines are loaded with active resistance, the value of which is commensurate with the modulus of the characteristic impedance of the line. Under these conditions, the SL attenuation monotonically increases with frequency. Physically, this phenomenon can be explained using an equivalent circuit.

It is valid if the line length does not exceed a quarter of the wavelength of the transmitted signal, i.e. with an electrically short line. The resistance of the line wires, together with the resistance formed by the resistances of active and capacitive leakages between the line wires, and the load resistance form a voltage divider. As the frequency increases, the modulus increases and the modulus decreases. Therefore, the transmission coefficient of this circuit decreases with increasing frequency, and the attenuation increases.

Additional amplitude-frequency distortions arise due to changes in the input resistance of the connecting line over the frequency range. Since the CL is the load of the LU, changes in the input resistance of the CL lead to a change in the voltage drop across the internal resistance of the broadcast signal source - the LU. But at a small value internal resistance LU these distortions are insignificant and are not taken into account.

To correct the frequency response of the SL, a special four-port network with lumped parameters is used - a correction circuit (CC). Its attenuation in the operating frequency range should change so that the total attenuation of SL and CC does not depend on frequency. The assumption that the total attenuation of the SL and CC is equal to the sum of the attenuations and is valid only in the case when the input impedance of the CC is constant in the operating frequency range and is equal to the load resistance. Otherwise, when connecting the CC to the trunk line, the load on the trunk line will change and its attenuation will change.

The CC should introduce the greatest attenuation at the lowest operating frequency. Up to frequencies of 500-700 Hz, the attenuation should remain approximately constant, and then smoothly decrease to zero at the highest operating frequency. The physical properties of SL and CC are different; line is a four-terminal network with distributed parameters, CC is a four-terminal network with lumped parameters. Therefore, it is impossible to achieve complete compensation of amplitude-frequency distortions introduced by SL using CC.

The more points on the frequency axis are taken for which the QC attenuation should coincide with the attenuation obtained from the idealized curve, the more complex the QC scheme is.

The CC must have a minimum number of customizable (selectable) elements. At the highest frequency, the CC attenuation should approach zero. Turning on the CC should not change the frequency characteristic of the attenuation of the link associated with it, in this case, the SL, otherwise frequency correction will turn into a complex and labor-intensive process of empirical selection of CC elements. When switching on a CC at the end of a line, a CC with a constant input impedance should be used, and when turning on at the beginning of a trunk, use a minimum output impedance. Reducing the output resistance of the CC is also desirable when switching on the CC at the end of the line, since this reduces the voltage of external noise induced on input circuit amplifier next after the CC. Constancy of the input impedance is also useful in cases where the CC is switched on before the LAN, since this stabilizes the LU mode.

Therefore, the CC must have a constant input impedance, minimal output impedance, minimal attenuation at the highest operating frequency and the smallest number of adjustable elements.

Basic QC schemes:

The simplest two-terminal network, connected in a circuit in series with the load or in parallel with the load, does not provide good correction, since the input impedance of such a CC depends on frequency and changes the frequency response of the line.

A full parallel circuit has a constant input impedance and a high output impedance that varies with frequency. A complete series circuit has a constant input impedance and a small output impedance that also varies with frequency. For this reason, a full series circuit is most suitable for SL correction. The T-bridge circuit provides a constant input impedance, but its output impedance is greater than that of a full series circuit. Therefore, it is less suitable for correcting diabetes, although it is found quite often in standard equipment.

The degree of complexity of two-terminal networks depends on the required correction accuracy. If the two-terminal networks and c contain two elements each, and are formed by a parallel connection of active resistance and capacitance, and a series connection of active resistance and inductance, then the calculated attenuation characteristic will coincide with the idealized one at two points - at (practically, in the region of lower frequencies) and at. If - are three-element, then a match is obtained at three points. With increasing requirements for the accuracy of frequency response correction, CC alone is not enough. Then two or more CCs are used, with additional CCs serving to correct the unevenness of the frequency response remaining after the introduction of the first CC.

Complicating QC for economic reasons is undesirable. Therefore, they are usually limited to the condition that the idealized and calculated CC attenuation curves coincide at three points, which are taken as one, and one intermediate one. The calculation formulas are significantly simplified if we take as an intermediate point the frequency at which the CC attenuation is equal to half the maximum.

Two-terminal circuits are synthesized based on the following considerations.

In the region of lower frequencies, resistances must be purely active. At the highest calculated frequency, it should vanish and approach infinity. This can be achieved by performing it in the form of a series and in the form of a parallel oscillatory circuit. The resonant frequencies of the circuits must be equal and coincide with the highest frequency of the operating range. The attenuation of the CC in the region of lower frequencies is determined by the relation and:

The steepness of the frequency response of the CC attenuation increases with increasing ratio, and accordingly the frequency of half attenuation increases. Losses in oscillatory circuits reduce the accuracy of correction at higher frequencies. Therefore, inductors should have the lowest active resistance possible. Capacitors must have low dielectric losses.

When recording gramophone records, to increase the signal-to-noise ratio, a rise in high frequencies is provided. And the electromagnetic pickup itself, as noted, gives an almost linear increase in EMF with frequency, starting from the very low frequencies. Because of this, to work with electromagnetic pickups, it is necessary to use corrector amplifiers with a normalized frequency response. Two sections of the frequency range are subject to correction. In the frequency range from 50 to 500 Hz, the gain should fall with a slope of 20 dB/decade. In the range from 500 to 2000 Hz it remains constant, and starting from a frequency of 2.12 kHz it should again decrease linearly. The frequency response curve is the inverse curve of the dependence of the oscillatory speed of the cutter during recording, which is standardized according to international standards.

So, three characteristic frequencies are visible on the frequency response, defining its appearance: 50, 500 and 2120 Hz. They correspond to time constants of 3180, 318 and 75 μs. They allow you to calculate corrective RC chains in the amplifier-corrector circuit. These circuits can be made in the form of passive correction circuits or in the form of correction elements included in a negative feedback circuit.

The need to introduce correction complicates the amplifier circuit. Usually a special correcting amplifier is used, which brings the signal from the output of the sound pickup to a level typical for other signal sources of the order of 0.15-0.3 V. Of course, given the low output voltage level of modern sound pickups, the amplifier must be with extremely low level of intrinsic noise and interference. Fans consider a tube correction amplifier to be the highest chic, although obtaining a low noise level from it is more than problematic.

The abbreviation RIAA, although belonging to the Recording Industry Association of America, since 1954 it has actually been associated throughout the world with the standard for correcting the frequency response of long-playing vinyl records, as opposed to the numerous standards that existed for older gramophone records, which were designed for a rotation speed of 78 revolutions per minute. Although the introduction of the standard developed by the Recording Industry Association of America (RIAA standard) was not welcomed in Europe, the introduction of a common international standard nevertheless it became the dictate of the time. The International Electrotechnical Commission, IEC, introduced a frequency equalization standard for long-playing vinyl records that turned out to be almost identical to the American standard. The only difference was that the IEC standard recommends cutting the lower audio frequencies in the recording playback mode, and, in order to reduce low-frequency rumble (the so-called rumble effect caused by the beating of the disk rotation speed), it is recommended to introduce attenuation with a level of -3 dB per frequency of 20 Hz (when translated into time characteristics, this corresponds to a time constant of 7950 μs). Most manufacturers of high-quality preamplifiers considered that their equipment would be equipped with electric turntables high quality, therefore, the problem of rumble will not concern them, which is why they ignored the IEC requirements. Consequently, the record equalization standard they used was actually the RIAA standard.
However, there is still often strong pressure on equipment manufacturers to change the parameters of players that comply with the RIAA standard by introducing correction of the amplitude-frequency response in the low-frequency region.

This policy is determined by the fact that:

• some tube power amplifiers turn out to be sensitive to saturation of the magnetic core of the output transformer in cases where a large amplitude signal is received at low frequencies (less than 50 Hz) (including from the rumble effect);
• Reflective woofers are very easily overloaded at frequencies below their cutoff frequencies due to too little damping caused by the movement of the cone. Reflective speakers mounted on baffle boards typically have a cutoff frequency just below 100 Hz, while free-standing reflective speakers have a cutoff frequency of 50 Hz, or even lower;
• Recordings on long-playing vinyl records are characterized by low-frequency (less than 20 Hz) noise due to deformations and vibrations of the player's disc.

Thus, from the above it follows that all these problems could be eliminated by introducing low-frequency correction in the playback stage of equipment that complies with RIAA standards.
One possible positive approach to this problem is the possible adoption of the IEC recommendations for a time constant of 7950 µs, but a more reasonable solution would be to introduce an appropriately designed high-pass filter having an attenuation at the edge of the range of about 12 dB per octave, or even greater, with a resonant frequency of about 10 Hz (the so-called resonant rumble filters for suppressing low-frequency noise, determined by the imperfection of the mechanical part of the player). The CD player somehow failed to identify the need to introduce a low-pass filter with a resonant frequency of 10 Hz to solve problems associated with poorly designed speakers or problematic output transformers. But then the question immediately arises, what does vinyl long-playing records have to do with it? Warping and rumble are purely mechanical problems, and therefore must be dealt with purely within these limits, and not with the use of electrical tricks.

In practical circuits, the operational amplifier is covered by negative feedback (NFB). Due to the phase shift between the input and output signal of the op-amp (as the frequency increases in a multistage amplifier, this phase shift increases) at some frequencies the feedback can become positive. If at these frequencies the gain of the amplifier is greater than unity, then self-oscillations occur at the output of the circuit. To eliminate the occurrence of these oscillations (self-excitation of the op-amp), frequency correction circuits are used.

Let's consider an amplifier covered by voltage feedback (Fig. 8.7). Let's assume that the circuit uses a three-stage op-amp. Let's determine its gain. We will consider the transfer characteristic of the op-amp to be ideal, i.e. U out =K O U input . Then

where  - feedback coefficient (depth). From here:

.

Here K=K 0 /(1+  TO 0 ) - closed-loop amplifier gain. If the value Co. is great, then

And
,

i.e. It practically does not depend on the gain of the op-amp.

Let us now turn to Fig. 8.7, b, c. Frequency is marked on the frequency axis f pr, at which the phase shift between the output and input signals reaches 180°. It is now easy to determine from the graph the presence of excitation conditions in the circuit. If the line K*= 1/ intersects the frequency response at a point corresponding to a frequency greater f pr, then false oscillations will occur in the circuit. In this case, the phase shift along the feedback circuit reaches a value greater than 360°. Consequently, the depth of the amplifier's negative feedback is limited by the stability condition of the op-amp. In Fig. 8.7, b the limits for changing the possible gain of the amplifier are indicated, at which the op-amp is not excited (area 1).

The most frequently used requirement in practice to ensure circuit stability, corresponding to the maximum possible phase margin in the OOS loop (with the accepted phase approximation at frequency f sr2- 90°, actually 45°), is as follows: straight K*= 1/ (dB) should intersect the frequency response segment with a slope of 20 dB/dec. In some cases, a smaller phase margin for self-excitation may be sufficient, so in amplifiers with negative feedback it is possible to use part of the section with a slope of 40 dB/dec.

E
If it is necessary to implement an amplifier with negative feedback, for which the formulated stability criterion is not satisfied, then frequency correction circuits must be introduced into the op-amp. The latter must, in the simplest case, change the frequency response of the op-amp so that the stability criterion for the required TO*. If the correction circuits are selected in such a way that the slope of the resulting op-amp frequency response is 20 dB/dec and it passes through the unity gain frequency point f T , then the amplifier has a fully corrected frequency response, which is called optimal.

Let's look at some correction chains. A corrective chain of a differentiating type (correction for phase advance) has become widespread (Fig. 8.8). The peculiarity of the frequency response of this circuit is its rise in the frequency range from f 4 to f 5 at a rate of 20 dB/dec.

Role R1 usually fulfills one of the internal resistances of the op-amp. Often and R1 implemented inside the op-amp. Therefore, correction of this type is reduced only to connecting a capacitor C1(sometimes R2) to the corresponding conclusions.

Corrective chain integrating type
a (correction for phase lag) is shown in Fig. 8.9. The Bode frequency response of this circuit in the frequency range from f 6 to f 7 falls at a rate of -20 dB/dec. The role of resistance R3 As a rule, the output resistance of the correction stage plays a role. Therefore, integrating-type correction in practice comes down to connecting a circuit R4С2.

TO
How are the considered chains used to correct two-stage amplifiers? In Fig. 8.10 shows the initial frequency response of a two-stage amplifier, frequency characteristics (curves 1 ’ , 2 ’ , 8 ’ ) used corrective chains (for them TO< 0) and the corresponding corrected frequency response (curves 1, 2, 3). The figure shows that a differentiating type correction chain makes it possible to perform both partial and optimal correction of the frequency response, in which the frequency response decline in the entire frequency band of the op amp is - 20 dB/dec (curve 2 in Fig. 8.10, b).

In practice, a number of other chains are used to correct the frequency response of op-amps. It is important to note that for each specific amplifier, the reference manuals recommend its own set of RC circuits connected to special pins (high-impedance points of the circuit). These points are selected in such a way that the values ​​of the elements of the correction circuit are small. The frequency response of modern two-stage amplifiers is corrected using one external correction circuit, while three-stage amplifiers are usually corrected using two circuits.

A number of op-amps have built-in frequency correction circuits, most often implemented on the basis of MOS capacitors, formed in the crystal simultaneously with other amplifier elements. Such amplifiers remain stable regardless of the amount of feedback, which is their undoubted advantage. (It simplifies the design of circuits based on them). However, internally compensated op amps have limited bandwidth and therefore do not take full advantage of the amplifier's dynamic properties for TO*>>1 (in them, frequency correction is performed for the worst case, i.e. for K*= 1).

We have learned to count acoustic design with a bass reflex and began to experimentally determine the dependence of the total electrical resistance of dynamic heads on frequency. Today we will try to understand the measurement results, after which we will consider methods for amplitude and frequency correction of emitters.

If you find impedance lows around 3 ohms, don't be discouraged. Some speaker models from well-known companies have dips of up to 2.6 Ohms, and sometimes even up to 2 Ohms! Of course, there is nothing good about this - the amplifiers overheat when working with such a load, especially at high volumes, and distortion increases.

For tube triode amplifiers, the minimums in the low frequencies and lower mids are especially dangerous. If the impedance here drops below 3 ohms, the end tubes may fail, but pentodes are not afraid of this.

It is important to remember that the output impedance of the amplifier is involved in setting the speaker filter. For example, if you make a 1 dB increase in the Fc region by connecting the speaker to a transistor amplifier with almost zero output resistance, then when working with a tube amplifier (typical value Rout = 2 Ohms) there will be no trace of afterburner. And the entire frequency response will be different. To get the same results, you will have to create a different filter.

A listener who never stops developing will eventually come to understand the value of good tube amplifiers. For this reason, I usually configure acoustics with a tube terminal, and when connecting to a transistor, I place a 10-watt non-inductive (no more than 4 - 8 mN) resistor with a resistance of 2 Ohms in series with the speaker.

If having transistor amplifier, you do not exclude the possibility of purchasing a lamp in the future, then when setting up and subsequent operation, connect your speakers through such resistors. When switching to lamps, you will not need to configure the speakers again; you just need to remove the resistors.

In the absence of a generator, a test CD with recording of test signals to evaluate the frequency response is suitable. In this case, you will not be able to smoothly change the frequency and, most likely, you will miss the very minimum impedance. Nevertheless, even an approximate estimate of the modulus Z will be useful, and for this, pseudonoise signals in one-third octave bands are even more convenient than sinusoidal ones. Such signals are on the test CD of the magazine “Salon AV” (No. 7/2002). As a last resort, you can avoid impedance measurements by limiting the return boost at the filter cutoff frequency to 1 dB. In this case, the impedance is unlikely to drop by more than 20%. For example, for a 4-ohm speaker this corresponds to a minimum of 3.2 ohms, which is acceptable.

Please note that you will have to “catch” the parameters of the filter elements necessary to correct the frequency response yourself. Preliminary calculation is needed so as not to miss “by a kilometer” from the beginning. Resistors are added to a simple filter of the low-pass/mid-range head for some manipulations with the frequency response, which may be required when tuning your speakers. If intermediate level If the sound pressure of this speaker is higher than the corresponding parameter of the HF head, it is necessary to connect a resistor in series with the speaker.

Inclusion options are shown in Fig. 6 a) and b).

The amount of required reduction in the output of the bass/midrange head, expressed in dB, will be denoted by N. Then:

where Rd is the average impedance of the speaker.

Instead of calculations, you can use Table 1.

Table 1

1 dB - = 10%, or level change by 1.1 times.

2 dB - = 25% - » - 1.25 times.

3 dB - = 40% - » - 1.4 times.

4 dB - = 60% - » - 1.6 times.

5 dB - = 80% - » - 1.8 times.

6 dB - = 100% - » - 2 times.

where Vс is the effective voltage value at the output of the amplifier. Vd - the same, on the dynamics. Vd is less than Vc due to the attenuation of the signal by resistor R1. In addition, N = NHF - NLF, where NLF and NHF are the sound pressure level developed, respectively, by the LF and HF heads.

These levels are averaged over the bands reproduced by the LF and HF heads. Naturally, NLF and NHF are measured in dB.

An example of a quick estimate of the required value of R1:

For N = 1 dB; R1 = Rd (1.1 - 1) = 0.1 Rd.

For N = 2 dB; R1 = Rd (1.25 - 1) = 0.25 Rd.

For N = 6 dB; R1 = Rd (2 - 1) = Rd.

More specific example:

Rd = 8 Ohm, N = 4 dB.

R1 = 8 ohms (1.6 - 1) = 4.8 ohms.

Let Rd be the rated power of the LF/MF loudspeaker, PR1 be the permissible power dissipated by R1.

You should not make it difficult to remove heat from R1, that is, you do not need to wrap it with electrical tape, fill it with hot glue, etc.

Features of preliminary calculation of the filter with R1.

For the circuit in Fig. 6 b) the values ​​of L1 and C1 are calculated for an imaginary speaker whose total resistance is: RS = R1 + Rd.

In this case, L1 is larger and C1 is smaller than the filter without R1.

For the circuit in Fig. 6 a) - the opposite is true: the introduction of R1 into the circuit requires a decrease in L1 and an increase in C1. It is easier to calculate the filter according to the diagram in Fig. 6 b). Use this exact scheme.

Additional correction of frequency response using a resistor.

If, to improve the frequency response uniformity, it is necessary to reduce the filter’s suppression of signals above the cutoff frequency, you can use the circuit shown in Fig. 7

R2 in this case gives a decrease in return in Fс. Above Fc, the output, on the contrary, increases compared to a filter without R2. If it is necessary to restore the frequency response close to the original one (measured without R2), L1 should be reduced and C1 increased in the same proportion. In practice, the range of R2 is within:

R2 = (0.1E1) i Rd.

Frequency response correction

The simplest case. In a fairly uniform characteristic, there is a zone of increased output (“presence”) in the mid-frequency region. You can use a corrector in the form of a resonant circuit (Fig. 8).

At resonance frequency

The circuit has a certain impedance value, in accordance with the value of which the signal at the speaker is attenuated.

Outside the resonance frequency, attenuation is reduced so that the circuit can selectively suppress presence.

It is convenient to use table 1a:

Example: it is necessary to suppress “presence” with a central frequency of 1600 Hz. Speaker impedance - 8 Ohms. Suppression level: 4 dB.

The specific shape of a loudspeaker's frequency response may require more complex correction.

Examples are in Fig. 9.

The case in Fig. 9 a) is the simplest. It is easy to select the parameters of the correction contour, since the “presence” has a “mirror” shape to the possible filter characteristic.

In Fig. 9 b) shows another possible option. It can be seen that the simplest circuit allows you to “exchange” one large “hump” into two small ones with a small dip in the frequency response in addition.

In such cases, you must first increase L2 and decrease C2. This will expand the suppression band to the required limits. Then you should bypass the circuit with resistor R3, as shown in Fig. 10. The value of R3 is selected based on the required degree of suppression of the signal supplied to the speaker in the band determined by the circuit parameters.

Fig.10

R3 = Rd (D - 1)

Example: you need to suppress the signal by 2 dB. Speaker - 8 Ohm. Refer to Table 1.

R3 = 8 ohms (1.25 - 1) = 2 ohms.

How the correction occurs in this case is shown in Fig. 9 c).

Modern loudspeakers are characterized by a combination of two problems: “presence” in the region of 1000 - 2000 Hz and some excess of the upper mids. Possible view The frequency response is shown in Fig. 11 a).

The correction method that is most free from harmful “side” effects requires a slight complication of the contour.

The corrector is shown in Fig. 12

The resonance of the L2, C2 circuit is needed, as usual, to suppress “presence”. Below Fp, the signal passes almost losslessly to the speaker via L2. Above Fp, the signal goes through C2 and is attenuated by resistor R4.

The corrector is optimized in several stages. Since the introduction of R4 weakens the resonance of the L2, C2 circuit, you should initially choose L2 more and C2 less. This will provide excess suppression on Fp, which is normalized after administration of R4.

R3 = Rd (D - 1), where D is the amount of suppression of signals above Fp.

D is selected according to the excess of the upper middle, referring to Table 1.

The stages of correction are roughly illustrated in Fig. 11 b).

In rare cases, a reverse effect on the slope of the frequency response using a correction circuit is required. It is clear that for this R4 must move to circuit L2.

The diagram is in Fig. 13.

The problematic frequency response and its correction for this case are shown in Fig. 14.

For a certain combination of L2, C2 and R4 values, the corrector may not have much suppression at Fp.

An example when just such a correction is necessary is shown in Fig. 15.

(To be continued)

3.2. High-frequency and low-frequency correction of the frequency response of a resistor amplifier

To adjust the frequency response real amplifier In order to bring it closer to the frequency response of an ideal amplifier (see Fig. 3.1), special correction circuits are used in the low-frequency and high-frequency regions.

The scheme for HF correction of the frequency response using corrective inductance Lk is shown in Fig. 3.8.

The operating principle of this circuit is based on an increase in the HF region of the resistance of the collector circuit (Rк + jwLк). Increasing this resistance with increasing w makes it possible to increase the RF gain of the cascade. A necessary condition for the effectiveness of this circuit is the high resistance of the external load resistance Rн >Rк. Otherwise, the low resistance Rн will shunt the collector circuit, while the gain of the cascade will be determined by the value of Rн and will depend little on Rк and Lк. The equivalent circuit of a cascade with HF correction at 1/Yi > Rн > Rк is shown in Fig. 3.9, from which it follows that the HF frequency response of the corrected amplifier is close to the frequency response of a parallel oscillatory circuit.

Consequently, if the parameters of the correction inductance Lk are not optimally selected, a rise may appear in the amplifier’s frequency response, causing distortion of the amplified signals. The frequency response and frequency response of an amplifier with RF correction at optimal and non-optimal parameters of the correction inductance Lk are shown in Fig. 3.10.

1. Lк< Lопт 2.Lк = Lопт 3.Lк >Lopt

It can be seen that the HF correction affects only the HF region (the region of short times - pulse fronts). When Lк > Lopt, the rise time is the shortest, however, an overshoot occurs on the output pulse signal.

The circuit for low-frequency correction of the amplifier's frequency response is shown in Fig. 3.11, where Rf and Cf are elements of low-frequency correction, which simultaneously serve as a low-frequency filter in the power circuit of transistor VT1.

The operating principle of the low-frequency correction circuit is based on increasing the resistance of the collector circuit in the low-frequency region, therefore, as in the inductive high-frequency correction circuit, this scheme effective only with high-resistance load Rн > Rк. The capacitance of the capacitor Cp is selected in such a way that at medium and high frequencies carried out 1/wСф<< Rф (то есть Сф шунтирует Rф), поэтому цепь Сф, Rф практически не оказывает влияния на работу усилителя на СЧ и ВЧ. На НЧ сопротивление Сф становится больше сопротивления Rф, это увеличивает сопротивление коллекторной цепи и как результат - понижает нижнюю граничную частоту полосы пропускания усилителя. При этом отношение Rф/Rк определяет максимально возможный подъем усиления с понижением частоты w, который однако, реально всегда бывает меньше по причине снижения усиления на НЧ из-за разделительного конденсатора Ср.

The frequency response and frequency response of the amplifier with optimal and non-optimal low-frequency correction parameters (1 - without correction, 2 - optimal correction, 3 - overcorrection) are shown in Fig. 3.12.

4. DESCRIPTION OF THE LABORATORY INSTALLATION.

The laboratory setup includes:

1) laboratory layout;

2) laboratory power supply;

3) universal voltmeter (type V7-15, V7-16).

4) generator of low-frequency signals (type G3-56, GZ-102).

Laboratory layout contains:

a) the investigated AC resistor amplifier with an emitter follower at the output to ensure high-resistance load of the amplifier (see Fig. 4.1.).

b) built-in pulse signal generator (with the ability to adjust the amplitude and duration of the pulses), located on the upper part of the laboratory model body.

The laboratory prototype is powered from a constant voltage source En = +12V. The appearance of the front panel with a schematic diagram of the laboratory layout applied to it is shown in Fig. 4.2.

5. OPERATION PROCEDURE

5.1. Study of the influence of the coupling capacitor on the characteristics of the amplifier.

a) Assemble the installation according to the diagram in Fig. 5.1. Place all switches in their original position 1.

Set the Uout value within 10...30 mV to ensure linear operation of the amplifier. By studying the dependence of Uout on the frequency f of the input signal (with a constant value of Uin), obtain and construct the frequency response of the amplifier at 2 values ​​of capacitance Cp (switch S4). When studying the frequency response, it is recommended to first evaluate the frequency region of uniform gain, where the number of samples can be reduced to 3...4. In the frequency ranges of frequency response changes (LF and HF), the number of sampling points should be increased to 4...5.

b) Connect a pulse signal from a rectangular pulse generator to the input of the amplifier under study (see section 4). Monitor the output voltage of the amplifier using an oscilloscope. Draw from the oscilloscope screen on one graph the shape of the pulses at the output of the amplifier (PA amplifier) ​​for two values ​​of Av.

Measure the magnitude of the decay of the flat part of the top of the pulse (in %) for two values ​​of Av.

Draw conclusions about the influence of the coupling capacitor Cp on the characteristics of the amplifier.

5.2. Study of the influence of collector resistance on amplifier characteristics.

Using the scheme and methods of clause 5.1. measure the nominal gain Ko, remove the amplifier's frequency response and PH for 2 values ​​of Rk. Plot the amplifier's frequency response and phase response for two values ​​of Rк.

Draw conclusions about the influence of collector resistance on the characteristics of the amplifier.

5.3. Study of the influence of low-frequency correction.

Set switch S4 to the position corresponding to the lower value of Av. Investigate the frequency response and frequency response of the amplifier for 3 values ​​of low-frequency correction parameters. Construct the amplifier's frequency response and frequency response for various low-frequency correction parameters.

Draw conclusions about the influence of Rph, Sph on the characteristics of the amplifier.

5.4. Study of the influence of HF correction

Set switch S1 to position Rк max, and switch S5 to position 1.

Investigate the frequency response and phase response of the amplifier for 3 values ​​of the correction inductance Lк. Construct the frequency response and frequency response of the amplifier for various parameters of inductive RF correction.

Draw conclusions about the influence of Lk on the characteristics of the amplifier.

5.5. Preparing a laboratory report.

The report must contain:

a) circuit of an alternating current resistor amplifier with low-frequency and high-frequency correction;

b) measurement results, tables and graphs required by laboratory tasks;

c) conclusion on the correspondence of the results obtained to the theoretical data.

6. CHECK QUESTIONS

1. Elements of temperature stabilization of the operating point of the transistor and their selection.

2. Operation of the resistor cascade in the low frequency region.

3. Operation of a resistor cascade in the HF region.

4. The influence of the decoupling capacitor Cp on the characteristics of the amplifier.

5. The influence of collector resistance Rк on the upper limit frequency and nominal gain.

6. The operating principle of inductive HF correction of a resistor amplifier.

7. Frequency response of the amplifier with optimal and non-optimal parameters of the HF correction elements.

8. PH of the amplifier with optimal and non-optimal parameters of the HF correction elements.

9. The principle of operation of low-frequency correction of a resistor amplifier.

10. Frequency response of the amplifier with optimal and non-optimal parameters of the low-frequency correction elements.

11. PH of the amplifier with optimal and non-optimal parameters of the low-frequency correction elements.

7. L I T E R A T U R A.

1. Ostapenko G. S. Amplifier devices. - M.: Radio and Communications, 1989, subsections 1.4, 1.5, 3.2, 4.8.

2. Voishvillo G.V. Amplifier devices. - M.: Radio and Communications, 1983, subsections 4.1.1, 4.7.3, 5.3.1, 5.3.3.

3. Mamonkin I. G. Amplifier devices. - M.: Communication, 1977, subsections 6.3, 7.3, 11.3.